Signal adjustment receiver circuitry

ABSTRACT

Systems and methods for adjusting a signal received from a communication path are disclosed. A receiver can receive a signal from a communication path which attenuates at least some frequency components of the signal. The receiver can include an equalization block that adjusts at least some of the frequency content of the received signal, a signal normalization block that provides a normalized signal amplitude and/or a normalized edge slope, and a control block. In one embodiment, the control block controls frequency adjustment in the equalization block for high frequencies but not for low frequencies. For low frequency adjustment, the control block controls the normalized signal amplitude in the signal normalization block. In this manner, controlled adjustment for low frequency content is performed in the signal normalization block.

BACKGROUND OF THE INVENTION

This invention relates to data communication, and more particularly to methods and apparatus for automatically adjusting the circuitry involved in such communication to compensate for losses in a data signal transmitted from transmitter circuitry to receiver circuitry.

Different signal transmission media tend to have different signal transmission characteristics. For example, a cable may have a different transmission characteristic than a printed circuit board backplane. In addition, each instance of any given type of transmission medium may have somewhat different characteristics, within a range that is typical for that type of transmission medium. It is also possible for a medium's transmission characteristics to change over time or as a result of environmental or operating factors.

Among the characteristics that can adversely affect the performance of a transmission medium are attenuation and phase shift. It is common for the amount of attenuation and phase shift to be frequency-dependent. Typically, both attenuation and phase shift tend to increase with increasing frequency. For convenience herein, attenuation, phase shift, and other forms of signal degradation are sometimes referred to generically as “losses.”

In order to have satisfactory transmission of a digital data signal, especially at high data rates or high frequencies, it may be necessary to compensate for losses in the signal being transmitted. Moreover, because such losses can vary from instance to instance and from time to time, it can be desirable for such compensation to be at least partly automatic or adaptive. A term that is often used for such compensation is equalization. The term pre-emphasis is also sometimes used for compensation or equalization that is performed at the transmitter, i.e., anticipating losses that will occur and compensating for them by modifying the signal before it is transmitted. When the term pre-emphasis is used, equalization may then be used as the term for compensation performed at the receiver.

Programmable circuitry such as programmable logic device (“PLD”) circuitry has capabilities that can be useful in supporting adaptive equalization. For example, a PLD or PLD circuitry may be one of the components involved in transmitting or receiving a signal needing adaptive equalization, or such circuitry may be used for controlling certain aspects of the circuitry that transmits and/or receives such a signal. Such programmable circuitry (e.g., PLD circuitry) can be especially useful in implementations of this invention because programmability aids in providing different parameters and/or procedures for addressing different transmission loss characteristics that may be encountered.

SUMMARY OF THE INVENTION

The disclosed invention is a technology for adjusting a digital signal received from a communication path. A communication path may attenuate some frequency components in a digital signal, and a receiver can include circuitry to compensate for the attenuation.

A receiver can include an equalization circuit that adjusts at least some of the frequency content of a digital signal, a signal normalization circuit that converts a signal amplitude to a normalized signal amplitude and converts a signal edge slope to a normalized edge slope, and a control circuit. As used herein, the term “slope” refers generally to a rate of rise or decline between “high” and “low” signal amplitude regions. The rise and decline can be linear or non-linear. The control circuit can control the frequency content adjustment in the equalization block and/or control the normalized signal amplitude or the normalized edge slope in the signal normalization circuit.

A receiver configuration having an equalization circuit and a signal normalization circuit provides flexibility. The signal normalization parameters in a signal normalization circuit can be changed to allow a receiver to operate in different modes and to interface with different devices. A signal normalization circuit, by providing adjustable signal normalization parameters, is also capable of adjusting frequency content, much like an equalization circuit. Therefore, a receiver can also have the flexibility of apportioning frequency adjustment operations between the equalization circuit and the signal normalization circuit.

The control block can include high pass filters and low pass filters having corner frequencies. In one embodiment, the corner frequency of the low pass filters can define a boundary frequency below which frequencies are characterized as “low frequencies,” and the corner frequency of the high pass filters can define a boundary frequency above which frequencies are characterized as “high frequencies.”

In one aspect of the invention, the signal normalization circuit can provide a normalized signal amplitude and a normalized edge slope that are separately controllable. A signal amplitude can be representative of a particular range of lower frequency content. In one embodiment, the control circuit can control the normalized signal amplitude in the signal normalization circuit to control low frequency content adjustment. Additionally, the control circuit may not control frequency adjustment in the equalization circuit for low frequencies. In this manner, controlled adjustment of low frequency content in a received signal can be performed in the signal normalization block and not in the equalization block.

In one aspect of the invention, a control circuit can control the signal normalization circuit based on an output of the equalization circuit. For example, a signal amplitude can be representative of a particular range of lower frequency content. The signal normalization circuit, by providing a normalized signal amplitude, can thereby adjust low frequency content at the output of the signal normalization circuit. In one embodiment, the control circuit can control the normalized signal amplitude in the signal normalization circuit to cause the low frequency content at the output of the signal normalization circuit to substantially equal the low frequency content at the output of the equalization circuit.

In one aspect of the invention, a control circuit can control the equalization circuit based on an output of the signal normalization circuit. For example, an edge slope can be representative of a particular range of higher frequency content. The signal normalization block, by converting signal edge slope to a normalized edge slope, can provide a particular high frequency content at the output of the signal normalization block. In one embodiment, the control circuit can control the equalization circuit so that the high frequency content at the output of the equalization circuit is substantially equal to the high frequency content at the output of the signal normalization circuit.

In accordance with one aspect of the disclosed technology, the control circuit can control the equalization circuit and the signal normalization circuit based on comparing the output of the equalization circuit with the output of the signal normalization circuit.

In one embodiment, the control circuit can measure the low frequency content at the output of the equalization and the low frequency content at the output of the signal normalization circuit using low pass filters. The control circuit can control the normalized signal amplitude in the signal normalization circuit to cause the low frequency contents to be substantially the same. As previously described herein, low frequency content can be defined based on a corner frequency of low pass filters in the control circuit.

In one aspect of the invention, the control circuit can measure the high frequency content at the output of the equalization and the high frequency content at the output of the signal normalization circuit using high pass filters. The control circuit can control the frequency adjustment in the equalization circuit to cause these high frequency contents to be substantially the same. As previous described herein, high frequency content can be defined based on a corner frequency of high pass filters in the control circuit.

In one aspect of the invention, the corner frequencies of the low pass filters and high pass filters in the control circuit can be predetermined or adjustable. In one embodiment, the corner frequency of the low pass filters can be adjusted based on the normalized signal amplitude of the signal normalization circuit and based on frequencies represented by the normalized signal amplitude. In one embodiment, the corner frequency of the high pass filters can be adjusted based on the normalized edge slope of the signal normalization circuit and based on frequencies represented by the normalized edge slope.

In one embodiment, the low pass filters and high pass filters can be formed by a serial connection of an adjustable resistance with an adjustable capacitance. The adjustable resistance can be a parallel, serial, and/or other arrangement of resistances and switches, and the adjustable capacitance can be a parallel, serial, and/or other arrangement of capacitances and switches. The switches can be controlled by a programmable logic device to adjust the resistances and capacitances in the filters to adjust their corner frequencies.

Further features of the invention, its nature and various advantages, will be more apparent from the accompanying drawings and the following detailed description.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an exemplary communication system in accordance with one aspect of the invention;

FIG. 2 is a diagram of exemplary component signals in a digital signal;

FIG. 3 is a graph diagram of exemplary attenuation and gain curves;

FIG. 4 is a circuit diagram of one embodiment of the signal normalization block of FIG. 1;

FIG. 5 is a circuit diagram of one embodiment of a stage in the signal normalization circuit of FIG. 4;

FIG. 6 is a block diagram of one exemplary control arrangement in the receiver of FIG. 1;

FIG. 7 is a circuit diagram of an exemplary low pass filter;

FIG. 8 is a circuit diagram of an exemplary high pass filter; and

FIG. 9 is a block diagram of one embodiment of a receiver that includes a programmable logic device.

DETAILED DESCRIPTION

The disclosed technology is an invention for adjusting a transmission signal at a receiver. Referring now to FIG. 1, there is shown a system 100 for communicating a signal between a transmitter 102 and a receiver 104. The communication path 106 between the transmitter 102 and the receiver 104 can include one or more communication media (e.g., wire, air, optical fiber) and/or one or more communication devices (e.g., router, hub, switch). At various points along the communication path 106, the communication path may include a digital or an analog signal. As used herein, a “digital signal” refers to a signal that represents information by using “high” and “low” signal amplitude regions and edge transitions between the two amplitude regions. Although the communication path 106 can include digital or analog signals at various points in the path, the receiver 104 of the disclosed technology receives a digital signal.

FIG. 2 shows a digital signal 202 having high and low amplitude levels and edge transitions between the two amplitude levels. Based on Fourier transforms, one skilled in the art will recognize that a digital signal 202 can include several component signals 204 that have different frequencies and amplitudes. The slope of edge transitions 206 in a digital signal can be indicative of high frequency content. As used herein, the term “slope” refers generally to a rate of rise or decline between “high” and “low” signal amplitude regions. The rise and decline can be linear or non-linear. An edge transition 206 having a greater slope can indicate a greater content of high frequency components than an edge transition having a lesser slope. The amplitude levels 208 of the digital signal can be indicative of low frequency content, such that a greater amplitude level can indicate a greater content of low frequency components than a lesser amplitude level. In one embodiment, the ranges of frequencies that are classified as “high frequencies” or “low frequencies” can be based on a boundary frequency, where high frequencies include those frequencies above the boundary frequency and low frequencies include those frequencies below the boundary frequency. In other embodiments, there can be more than one boundary frequency. For example, in one embodiment, high frequencies can include frequencies that are above a first boundary frequency, and low frequencies can include frequencies that are below a second boundary frequency. The boundary frequency or frequencies can be constant or adjustable. Boundary frequencies will be described in more detail herein in connection with FIG. 7.

Referring now to FIG. 3, and with continuing reference to FIG. 1, exemplary graphs of attenuation and gain are shown. Attenuation as used herein refers to decreases in signal energy that can occur because of various phenomena in or affecting a communication system or because of filters or other circuitry or otherwise. Gain, on the other hand, refers to increases in signal energy. A signal in a communication system may experience attenuation in a communication path 106 (FIG. 1). One skilled in the art will recognize that component signals (e.g., 204, FIG. 2) that have higher frequency typically experience greater attenuation than component signals that have lower frequency, as illustrated by the exemplary attenuation graph 302 in FIG. 3. As higher frequency components in a signal become attenuated, a signal's edge transitions (e.g., 206, FIG. 2) can decrease in slope, which may undesirably affect circuit timing and/or other circuit operations.

A purpose of an equalization block 108 (FIG. 1) in a receiver is to provide gain to offset attenuation that may exist in a communication path 106. Because the attenuation effects of a communication path 106 may change over time, the equalization block 108 may need to make adjustments to adapt to the changes. The gain graph 304 in FIG. 3 shows exemplary gain curves that an equalization block 108 can provide. The determination of which gain curve best offsets the attenuation in a communication path 106 can be made by a control block 112 (FIG. 1), which will be described later herein in connection with FIG. 6. FIG. 3 is exemplary and does not limit the number, shape, and/or magnitudes of attenuation and gain curves within the scope of the invention.

Referring again to FIG. 1 and in accordance with one aspect of the disclosed invention, the illustrated receiver 104 includes an equalization block 108, a signal normalization block 110, and a control block 112. A receiver having an equalization block 108 and a signal normalization block 110 provides flexibility. The signal normalization parameters in a signal normalization block 110 can be changed to allow the receiver 104 to operate in different modes and data rates and to interface with different devices. A signal normalization block 110, by providing adjustable signal normalization parameters, is also capable of adjusting frequency content, much like an equalization block 108. Therefore, a receiver 104 can also have the flexibility of apportioning frequency adjustment operations between the equalization block 108 and the signal normalization block 110.

In general, the equalization block 108 can provide gain and/or attenuation and can adjust high frequency content specifically 118 and/or adjust frequency content in general 120. The equalization block can optionally provide low frequency content adjustment separately (not shown). Frequencies can be characterized as “high frequencies” or “low frequencies” based on one or more boundary frequencies, which will be described in more detail in connection with FIG. 6. In one embodiment, the equalization block 108 can adjust the frequency content of a digital signal received from the communication path 106 according to a gain curve (e.g., 304, FIG. 3) to provide a resulting post-equalization signal. One skilled in the art will recognize that various existing technologies for adjusting frequency content can be used in the equalization block 108. The equalization block 108 can be controlled by the control block 112, which will be described later herein in connection with FIG. 6.

In accordance with one aspect of the invention, a post-equalization signal is communicated to the signal normalization block 110, which can convert the amplitude and/or the edge slope in the post-equalization signal to a normalized amplitude and/or a normalized edge slope, respectively. In one embodiment, the signal normalization block 110 can provide a normalized signal amplitude 114 and/or a normalized edge slope 116 based on industry standards, government regulations, and/or other requirements. Generally, because a receiver 104 and a transmitter 102 are designed to work together, the receiver 104 can know the signal amplitudes (e.g., 208, FIG. 2) and slope of edge transitions (e.g., 206, FIG. 2) used by a transmitter 102. Therefore, the signal normalization block 110 in the receiver 104 can provide a normalized signal amplitude 114 and/or a normalized edge slope 116 that is the same as those used by the transmitter 102 for providing a digital transmission signal.

In one embodiment, the signal normalization block 110 can provide a normalized signal amplitude 114 and a normalized signal edge slope 116 based on the operating requirements of a particular application. For example, a digital signal received by a receiver 104 may be intended for use by multiple devices that have different voltage requirements. One device may require a normalized signal amplitude of 600 mV, for example, and another device may require a normalized signal amplitude of 1200 mV, for example. Depending on which device may be interacting with the receiver 104, the control circuit 112 can change the normalized signal amplitude 114 accordingly. As another example, a transmitter 102 and a receiver 104 may have different modes of operation that use different transmission frequencies. If a digital signal communicated between the transmitter 102 and receiver 104 has a signal frequency of one megabit/second, a normalized edge slope of one nanosecond may be appropriate. However, if the digital signal has a signal frequency of ten gigabits/second, a normalized edge slope of one nanosecond may be too slow and the control circuit 112 may need to increase the normalized edge slope 116. Accordingly, in some embodiments, the normalized signal amplitude 114 and/or the normalized edge slope 116 in the signal normalization block 110 can be adjustable and can be controlled by a control block 112.

FIG. 4 shows a circuit embodiment of the signal normalization block of FIG. 1. The illustrated signal normalization circuit 400 includes at least two stages 402, 404. The first stage 402 has connections INP1 406 and INN1 408 for receiving a post-equalization signal. If the post-equalization signal is a differential signal, then both the INP1 and INN1 connections 406, 408 can be used to receive the post-equalization signal. If the post-equalization is a single-ended signal, then the INP1 connection 406 can be used to receive the post-equalization signal while the INN1 connection 408 can be connected to a reference voltage, or vice versa. For ease of explanation, it will be assumed that the post-equalization signal is a differential signal that can have a “high” value and a “low” value. When the post-equalization signal is “high,” the INP1 signal 406 is greater than the INN1 signal 408, and when the post-equalization is “low,” the INP1 signal 406 is less than the INN1 signal 408.

The INP1 connection 406 is connected to the gate of an INP1 field effect transistor (FET) 410, and the INN1 connection 408 is connected to the gate of an INN1 FET 412. It will be assumed herein that when the differential signal is “high,” the INP1 FET 410 will be ON and the INN1 FET 412 will be OFF, and when the differential signal is “low,” the INP1 FET 410 will be OFF and the INN1 FET 412 will be ON. The source node of the INP1 FET and the source node of the INN1 FET are connected to each other 414, and both are connected to one node of a variable current source 416. The other node of the variable current source is connected to a low reference voltage 418. The drain node of the INP1 FET 410 is connected to one node of an INP1 variable resistance 420, and the other node of the INP1 variable resistance 420 is connected to a high reference voltage 422. Similarly, the drain node of the INN1 FET 412 is connected to one node of an INN1 variable resistance 424, and the other node of the INN1 variable resistance 424 is connected to the high reference voltage 422.

The drain node of the INP1 FET 410 will be referred to herein as OUTP1 426, and the drain node of the INN1 FET 412 will be referred to herein as OUTN1 428. The voltage difference between the OUTP1 and OUTN1 nodes 426, 428 is a differential signal that is the output of the first stage 402. The output differential signal can be defined by either (OUTP1−OUTN1) or (OUTN1−OUTP1). For ease of explanation, it will be assumed from this point on that the output of the first stage is defined by (OUTP1−OUTN1). Additionally, OUTP1 426 is connected to the gate node of an INP2 FET 430 in the second stage 404, and OUTN1 428 is connected to the gate node of an INN2 FET 432 in the second stage 404.

Referring to the first stage 402, when the post-equalization signal has a “high” value, the INP1 FET 410 is ON and the INN1 FET 412 is OFF. When the INN1 FET 412 is OFF, OUTN1 428 has the same voltage as the high reference voltage V_(H) 422. When the INP1 FET 410 is ON and the INN1 FET 412 is OFF, the current I established by the variable current source 416 flows through the INP1 FET 410 and the INP1 variable resistance R_(INP1) 420. Thus, the voltage at OUTP1 426 is (V_(H)−I*R_(INP1)) . Accordingly, when the post-equalization has a “high” value, the output of the first stage is (OUTP1−OUTN1)=(V_(H)−I*R_(INP1))−V_(H)=−I*R_(INP1). When the post-equalization has a “low” value, the INP1 FET 410 is OFF and the INN1 FET 412 is ON. When the INP1 FET 410 is OFF, OUTP1 426 has the same voltage as the high reference voltage V_(H) 422. When the INN1 FET 412 is ON and the INP1 FET 410 is OFF, the current I established by the variable current source 416 flows through the INN1 FET 412 and the INN1 variable resistance R_(INN1) 424. Thus, the voltage at OUTN1 428 is (V_(H)−I*R_(INN1)). Accordingly, when the post-equalization has a “low” value, the output of the first stage is (OUTP1−OUTN1)=V_(H)−(V_(H)−I*R_(INN1))=+I*R_(INN1). By this operation, it can be seen that as the post-equalization signal varies between “high” and “low” values, the output of the first stage 402 will vary between −I*R_(INP1) and +I*R_(INN1), which are normalized signal amplitudes. Therefore, in the illustrated embodiment of FIG. 4, the signal normalization circuit 400 can adjust the normalized signal amplitude (114, FIG. 1) by changing the amount of current I established by the variable current source 416 and/or by changing the resistances R_(INP1) and R_(INN1) of the variable resistances 420, 424.

The output nodes OUTP1 and OUTN1 426, 428 are connected to the gate nodes of FETs 430, 432 in the second stage 404. Because the gate nodes of these FETs 430, 432 have gate capacitances, the variable resistances 420, 424 in the first stage 402 and the gate capacitances in the second stage 404 together form RC circuits that have RC time constants. The RC circuits cause the voltages at nodes OUTP1 and OUTN1 426, 428 to rise and decline exponentially based on the RC time constants, thereby defining a normalized edge slope in the output signal (OUTP1−OUTN1). The RC time constants can be adjusted by changing the resistances R_(INP1) and R_(INN1) 420, 424. This adjustment changes the slope of edge transitions in OUTP1 426 and OUTN1 428, and, as a consequence, the slope of edge transitions in the output signal (OUTP1−OUTN1) is also adjusted. Therefore, in the illustrated embodiment of FIG. 4, the signal normalization circuit 400 can provide a particular normalized edge slope (116, FIG. 1) by adjusting the resistances R_(INP1) and R_(INN1) of the variable resistances 420, 424.

In the illustrated embodiment of FIG. 4, the second stage 404 is the same as the first stage 402. The second stage 404 receives a differential signal via the input nodes INP2 426 and INN2 428 and provides a differential output signal (OUTP2−OUTN2). The output of the second stage 404 can be connected to the input of yet another stage or can be connected to the output of the signal normalization circuit.

FIG. 5 shows an embodiment of one stage of the signal normalization circuit of FIG. 4 in which circuit implementations of a variable resistance and a variable current source are shown. In general, a variable resistance can be implemented by a serial, parallel, and/or other arrangement of switches and resistances. FIG. 5 shows a parallel arrangement of resistances 502 in which each branch of the parallel arrangement can be included or excluded by closing or opening a switch 504. In one embodiment, the resistances 506 in each branch can have substantially the same values. In another embodiment, two or more resistances 506 in the arrangement 502 can have substantially different values. Although the switches 504 are illustrated as p-channel field effect transistors, other types of devices or circuits can be used to provide switching capability. The illustrated variable resistance circuit 502 does not limit the scope of the invention and other arrangements and/or types of switches and resistances can be used. The switches 504 can be operated by a programmable logic device (“PLD”), which will be described later herein in connection with FIG. 9.

In the illustrated embodiment of FIG. 5, the variable current source 416 of FIG. 4 is implemented using a current mirror circuit. Although a current mirror circuit and its operation will be understood by one skilled in the art, a brief description is provided here. In the current mirror circuit, a reference current can be established to flow through a reference-side FET 508. The gate and drain nodes of the reference-side FET 508 can be connected to each other, and the gate node of the reference-side FET 508 is also connected to the gate node of a “mirroring” FET 510. By this configuration, a mirror current can be established to flow through the mirroring FET 510. The amount of mirror current flowing through the mirroring FET 510 may or may not be the same as the amount of reference current flowing through the reference-side FET 508. The relationship between a mirror current and a reference current can depend upon the sizes of the reference-side FET 508 and the mirroring FET 510, among other things. When the reference current changes, the mirror current changes correspondingly based on the relationship. Each stage (e.g., 402, 404, FIG. 4) of the signal normalization circuit 400 of FIG. 4 can include a mirroring FET whose gate node is connected to the gate node of the reference-side FET 508. The size of each mirroring FET (e.g., 510) can determine the relationship of the mirror current in each stage to the reference current.

In the illustrated embodiment of FIG. 5, the reference current can be established by using a variable resistance 512. As previously described herein, a variable resistance 512 can be implemented by an arrangement of resistances 506 and switches 504. The switches 504 in the variable resistance 512 can be controlled by a programmable logic device (not shown). The illustrated implementation of a variable current source is exemplary and does not limit the scope of the invention. The current mirror circuit can include bi-junction transistors rather than field effect transistors. In another embodiment, a circuit other than a current mirror can be used to provide a current source.

Referring again to FIG. 1, the operation of the control block 112 in the receiver 104 will now be described. A control block 112 in the receiver 104 can control various operations in the equalization block 108 and in the signal normalization block 110. In general, the equalization block 108 can provide gain and/or attenuation and can adjust high frequency content specifically 118 and/or can adjust frequency content generally 120. The control block 112 can control the amount of gain or attenuation provided by the equalization block 108. In one embodiment, the control block 112 can control the general adjustment of frequency content 120 in equalization block 108. In one embodiment, the control block 112 can control only high frequency adjustment 118 in the equalization block 108. The equalization block can optionally provide low frequency content adjustment separately (not shown). The control block 112 may or may not control this low frequency content adjustment. The signal normalization block 110 can in general provide a normalized signal amplitude 114 and/or a normalized edge slope 116. In one embodiment, the control block 112 can control both the normalized signal amplitude 114 and the normalized edge slope 116. In one embodiment, the control block 112 can control only the normalized signal amplitude 114 or only the normalized edge slope 116, but not both.

As shown in FIG. 1, the control block 112 can receive a post-equalization signal at the output of the equalization block 108 and a normalized signal at the output of the signal normalization block 110. The control block 112 can control operations in the equalization block 108 and in the signal normalization block 110 based on these output signals. In this manner, the equalization block 108, the signal normalization block 110, and the control block 112 together form a feedback loop. The operation of the feedback loop can vary depending on the particular device or application in which the receiver 104 is used. One example is shown in FIG. 6.

FIG. 6 shows one embodiment of the receiver 104 of FIG. 1 in which controlled low frequency adjustment is provided in the signal normalization block. In the equalization block 602, the high frequency adjustment 604 and/or a general frequency content adjustment 608 can be controlled by the control block 606. If the equalization block 602 provides separate low frequency adjustment (not shown), the separate low frequency adjustment is not controlled by the control block 606. In the signal normalization block 608, the normalized signal amplitude 610 is controlled by the control block 606. The signal normalization block 608 can provide a normalized edge slope 612 that may or may not be controlled by the control block 606. In this configuration, controlled low frequency adjustment is provided in the signal normalization block 608 (by way of the normalized signal amplitude 610), and separate low frequency adjustment in the equalization block, if any, is not controlled. Benefits of using this configuration include, but are not limited to, simplifying the operation of the equalization block 602 and/or achieving greater gain for high frequency component signals relative to low frequency component signals in the equalization block 602.

In the illustrated embodiment, the control block 606 includes high and low pass filters 614-620. In one embodiment, a corner frequency for a low pass filter 614, 616 can be a frequency below which the low pass filter provides substantially no attenuation and above which the low pass filter provides attenuation. Similarly, a corner frequency for a high pass filter 618, 620 can be a frequency above which the high pass filter provides substantially no attenuation and below which the high pass filter provides attenuation. Attenuation can occur at different rates, such as a rate of decline of 20 dB/decade, for example, or a rate of decline of more than 20 dB/decade. The corner frequencies for a high pass filter 618, 620 and a low pass filter 614, 616 can be the same or substantially the same frequency or can be substantially different frequencies. The corner frequencies of the high and low pass filters 614-620 can be constant or adjustable.

In one embodiment, a corner frequency can correspond to a boundary frequency that defines which frequencies are categorized as “high frequencies” and which frequencies are categorized as “low frequencies.” For example, a corner frequency for a low pass filter 614, 616 can be the frequency below which frequencies are characterized as “low frequencies.” Similarly, a corner frequency for a high pass filter 618, 620 can be the frequency above which frequencies are characterized as “high frequencies.”

In one embodiment, corner frequencies for the high and low pass filters 614-620 can be predetermined and may not be changed or may not be adjustable. In one embodiment, the corner frequencies for the high and low pass filters 614-620 can be adjustable and can, for example, be adjusted based on a normalized signal amplitude 610 or a normalized edge slope 612. As previously described herein, a normalized signal amplitude 610 can be representative of a particular range of lower frequencies, and a normalized edge slope 612 of a signal normalization block 608 can represent a particular range of higher frequencies. In one embodiment, the corner frequency for a low pass filter 614, 616 can be adjusted based on a normalized signal amplitude 610. The corner frequency of a low pass filter 614, 616 can be chosen so that a substantial portion of the frequency content represented by the normalized signal amplitude 610 is below the corner frequency. In this manner, a normalized signal amplitude 610 can determine the range of frequencies that is categorized as “low frequencies.” In one embodiment, the corner frequency for a high pass filter 618, 620 can be based on a normalized edge slope 612. For example, the corner frequency of a high pass filter 618, 620 can be adjusted so that a substantial portion of the frequency content represented by the normalized edge slope 612 is above the corner frequency. In this manner, a normalized edge slope 612 can determine the range of frequencies that is categorized as “high frequencies.” Adjustable corner frequencies will be described in more detail herein in connection with FIGS. 7-9.

In the illustrated embodiment of FIG. 6, the control block 606 includes a first low pass filter 614 that provides the low frequency content of a post-equalization signal and a second low pass filter 616 that provides the low frequency content of a normalized signal. A low frequency content comparator 622 compares the outputs of the two low pass filters 614, 616. In one embodiment, the low frequency content comparator 622 can measure the energy levels of the outputs of the low pass filters 614, 616 and can adjust the normalized signal amplitude 610 in the signal normalization block 608 in a way that substantially equalizes the energy levels. By this operation, the normalized signal amplitude 610 of the signal normalization block 608 can be controlled to substantially equalize the low frequency content of the normalized signal and the low frequency content of the post-equalization signal. Therefore, this operation may assume that low frequency content is not attenuated by a communication path (106, FIG. 1) or is only attenuated in a manner that does not necessitate compensation.

The illustrated control block 606 also includes a first high pass filter 618 that provide the high frequency content of a post-equalization signal and a second high pass filter 620 that provides the high frequency content of a normalized signal. A high frequency content comparator 624 compares the outputs of the two high pass filters 618, 620. In one embodiment, the high frequency content comparator 624 can measure the energy levels of the outputs of the high pass filters 618, 620 and can control the equalization block 602 in a way that substantially equalizes the energy levels. By this operation, the frequency content adjustment 608 and/or high frequency adjustment 604 in the equalization block 602 can be adjusted to substantially equalize the high frequency content in the post-equalization signal and the high frequency content in the normalized signal.

A comparator 622, 624 in the control block 606 can be implemented in various ways. In one embodiment, a comparator 622, 624 can include circuitry for measuring the energy level of a signal. A comparator 622, 624 can include circuitry for rectifying the outputs of the filters. The rectified signals can be compared by an error amplifier, such as an operational amplifier circuit, and the error amplifier can produce a control signal based on the comparison. In one embodiment, the error amplifier can produce an analog control signal. In one embodiment, the analog control signal can be converted into a digital signal by an analog-to-digital converter, or the error amplifier can produce a digital control signal. In one embodiment, the digital signal can be communicated to a programmable logic device (PLD). Use of a PLD for control operations will be described herein in connection with FIG. 9. The embodiments described herein are exemplary and are not limiting. One skilled in the art will recognize that various circuits exist for implementing the comparator capabilities described herein.

Referring now to FIG. 7, there is shown one embodiment of a low pass filter 700 that includes an adjustable resistance 702 connected to an adjustable capacitance 704. Similarly, FIG. 8 shows one embodiment of a high pass filter 800 that includes an adjustable capacitance 802 connected to an adjustable resistance 804. An adjustable resistance 702, 804 can be implemented using a parallel, series, and/or other arrangement of resistances and switches, and an adjustable capacitance 704, 802 can be implemented using a parallel, series, and/or other arrangement of capacitances and switches. The switches can be opened or closed by a programmable logic device (not shown) to provide a desired resistance and/or a desired capacitance. By adjusting the resistance and/or capacitance, the corner frequencies of the filters 700, 800 in FIGS. 7-8 can be changed.

FIG. 9 shows one embodiment of the control block 112 of FIG. 1 (and also control block 606 of FIG. 6) in which a programmable logic device (PLD) 902 can be used to perform control operations. A PLD 902 can be connected to switches in an adjustable resistance or an adjustable capacitance, such as those shown in FIG. 5 and in FIGS. 7-8, and can be programmed to adjust the resistances and/or capacitances. For example, in connection with FIG. 5, the PLD 902 can be programmed to adjust the resistances 502 connected to output nodes OUTP1 and OUTN1 to provide a normalized edge slope. The PLD can also be programmed to adjust the resistance 512 in the current mirror to provide a normalized signal amplitude. As another example, in connection with FIGS. 7-8, the PLD 802 can be programmed to adjust the resistances 702, 804 and/or capacitances 704, 802 in the filters 700, 800 to provide desired corner frequencies. In one embodiment, when a PLD 902 changes the normalized edge slope 904 and/or the normalized signal amplitude 906 in the signal normalization block 908, the PLD 902 can also change the corner frequencies of the high and low pass filters 910, 912 in the control block accordingly. The PLD 902 or another component in the control block 916 can control the frequency content adjustment in the equalization block 914 based on the corner frequencies of the filters 910, 912 in the control block 916.

Although the PLD 902 of FIG. 9 is shown to be part of the control block 916, the control block 916 may only utilize a portion of the PLD resources so that the PLD 902 can be used for other purposes as well. Accordingly, the PLD 902 need not be physically part of the control block 916. FIG. 9 is exemplary and other arrangements and numbers of circuit components can be used to provide the capabilities described herein.

Accordingly, what has been describe thus far are systems and methods for adjusting a signal received from a communication path. A receiver can receive a signal from a communication path that attenuates at least some frequency components of the signal. The receiver can include an equalization block that adjusts at least some of the frequency content of the received signal, a signal normalization block that provides a normalized signal amplitude and/or a normalized edge slope, and a control block. The control block can control the frequency content adjustment in the equalization block and/or the normalized signal amplitude or the normalized edge slope in the signal normalization block. In one embodiment, the control block controls frequency adjustment in the equalization block for high frequencies but not for low frequencies. For low frequency adjustment, the control block controls the normalized signal amplitude in the signal normalization block. In this manner, controlled adjustment for low frequency content is performed in the signal normalization block. One skilled in the art will appreciate that any embodiment described and/or illustrated herein is exemplary and does not limit the scope of the invention as defined by the following claims. 

1-16. (canceled)
 17. A signal normalization circuit for providing a normalized signal amplitude and a normalized edge slope, comprising: a first stage circuit comprising: an adjustable current source, a first field effect transistor having a source node coupled to the adjustable current source, a gate node for receiving an input signal, and a drain node; a second field effect transistor having a source node coupled to the adjustable current source, a gate node, and a drain node, wherein the gate node is coupled to one of: the input signal and a reference voltage; a first variable resistance connected to the drain node of the first field effect transistor, and a second variable resistance connected to the drain node of the second field effect transistor; and a second stage circuit comprising: a third field effect transistor having a gate node connected to the drain node of the first field effect transistor, and a fourth field effect transistor having a gate node connected to the drain node of the second field effect transistor.
 18. The signal normalization circuit of claim 17, wherein each of the first variable resistance and the second variable resistance comprises an arrangement of switches and resistances.
 19. The signal normalization circuit of claim 18, further comprising a programmable logic device coupled to the switches, wherein the programmable logic device is programmed to configure the switches to provide a normalized edge slope.
 20. The signal normalization circuit of claim 17, wherein the adjustable current source comprises a current mirror circuit having a variable resistance for providing an adjustable current.
 21. The signal normalization circuit of claim 20, wherein the variable resistance of the current mirror circuit comprises an arrangement of switches and resistances.
 22. The signal normalization circuit of claim 21, further comprising a programmable logic device connected to the switches, wherein the programmable logic device is programmed to configure the switches to provide a normalized signal amplitude. 